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Click on the following bookmarks for further details of a Transient Tool for emc
reduction by series inductor
Charged capacitor discharge
Suppression filter step response
Analysis of hybrid transient
Analysis of the thermal
noise in a resistor
Transient voltage clamping
The voltage waveform utilised by this tool has a linearly rising edge followed by
an exponential decay, and a defined source resistance.
The circuit arrangement allows the above source voltage to be applied
to a resistive load through a series resistance.. The load can be shunted with a voltage
clamp device, which is open circuit until its threshold voltage is reached. At threshold,
it maintains a constant voltage across itself while the source voltage is capable of
maintaining the threshold. The clamp also has two series elements, an inductor and
resistor. These can be used to make realistic estimates of the effect of lead inductance
and pcb track impedances.
The data entry form is shown below:
The circuit arrangement is shown below:
The analysis provides total energy, peak power, peak current and
peak voltage in the load and series resistor. The clamp analysis provides total energy,
peak power and peak current. The load voltage can be plotted, and an example plot is shown
The load voltage when a 1500V peak voltage with a 500ohm load is
clamped by a 36V clamping device. A poor ground with 50mohm resistance raises the peak
voltage to 50V. The rise time of the pulse is 1us, and the decay time 5us.
voltage reduction by series inductor
The waveform shown on the data entry form below covers many requirements, consisting of
a linearly rising edge, followed by an exponential decay.
The waveform is defined by the tool in terms of the risetime to 50%
of the peak value, and the time from the pulse onset until decay to 50% of the peak.
During analysis, the time constant of the decay is displayed for information. The circuit
arrangement used by the inductor transient attenuator is shown below:
A spot analysis is produced of peak current, voltage and power in
the series resistor and load resistor, and also of total energy. Peak voltage and current
at the inductor are also presented. A plot can be produced, which graphs the load voltage
and inductor voltage against time.The performance of an inductive transient attenuator
varies for a given inductance according to the pulse shape.
For instance, an analysis of a 500V peak transient, of 8us rise and
20us decay times, shows a peak voltage of 48V at the load. Increasing the decay time to
200us, increases the peak load voltage to 86V, and also increases the total energy
delivered to the load from 21.5mJ to 141mJ. The plot above shows the 200us decay time
pulse, with the blue curve showing the inductor voltage, and the black curve the load
The charged capacitor discharge tool uses the circuit arrangement shown on the form
The capacitor C1 holds the initial charge, and the resistor Rs can
represent both source resistance, and any additional series resistance the user may wish
to add. Rl represents the load resistance, and C2 the charge balancing capacitor.
In the example figures shown on the form above, an ESD-like test has
been simulated. A 1500V charge on a 150pF capacitor has been discharged through 150 ohms
into a 50 ohm load. Without the charge balancing capacitor C2, the peak voltage at the
load would have been:
VPeak = (1500 * 50) / (50+150) = 375 volts.
Use of the 1nF capacitance has reduced the peak to 109 volts, and a
larger capacitance would reduce it further. The example plot below shows the same
arrangement as above, but with C2 increased to 10nF.
The peak voltage has been further reduced to less than 20 volts,
albeit at the expense of increased capacitance on the line.
EMI Suppression Filters consist of shunt capacitances and/or series
inductances. Their general response to the application of a step voltage is to produce a
delayed rising edge at a resistive load.
This effect can be of particular importance when the filtered line is
carrying a fast digital signal. If the signal risetime at the load exceeds the period of
the signal, the peak signal voltage is not reached, and the signal is effectively
A second effect occurs when a threshold voltage is important; the
threshold is delayed by the filter, and component tolerances in the filter may cause
different signal lines to produce staggered thresholds.
The tool allows six different filter circuits to be selected, which can
be shown on-screen via the Diagram button. The circuits are Capacitor, Inductor, L-C
Filter, C-L Filter, Pi Filter and T Filter. Source and Load Resistance can be specified,
and Inductors can also have a series resistance. The Filter Step form is shown below:
Application of the step then shows the load response; expand the timescale until the
rising edge can be seen. Comparison of the edge with the digital signal period will show
if the signal could suffer attenuation. Note that inductor/capacitor combinations can
cause ringing when an edge is applied, especially when the load resistance is high. A spot
analysis at a defined time is also available.
The diagram belows shows the step response of a pi filter, of 50nF
capacitance per section, and 1mH inductance, to a 15V step input. The source resistance is
50 ohms, and the load resistance is 50k Ohms.
hybrid transient protector
High energy voltage transients, originating from lightning strikes
for instance, produce two problems for single suppression devices. A silicon transient
suppression diode, or metal oxide varistor switches rapidly, but may have peak voltage and
total energy ratings below those produced by the transient. A gas discharge tube however
has very much higher peak current capability, but also has a finite switching time. This
results in a significant voltage overshoot at the load, which can be damaging.
A so-called 'hybrid' solution, combining gas discharge tube and diode
eliminates the overshoot and has high energy handling. However, a series impedance
(sometimes called a co-ordinating element) is required to produce a sufficiently high
voltage to allow the gas tube to switch on. The tool allows both a series resistance and
inductance to be specified. An inductor allows a lower low frequency circuit impedance,
but its voltage drop is dependant upon the rate of voltage increase. At high transient
dV/dt, the inductor voltage drop can be sufficient to prevent the diode from switching on.
The circuit arrangement analysed by the tool is shown below.
In addition to specification of the basic circuit values, the gas
discharge tube characteristics can be set, including the voltage overshoot against dV/dt.
This data is accessed from a separate form called from the hybrid protection tool form,
which is itself shown below.
The voltage overshoot increases with the rate of voltage increase;
this increase is logarithmic however, which means that the actual switching time decreases
with decreasing risetime. A voltage overshoot figure can be set for each decade of
risetime rate between 102 V/s to 1012 V/s. A typical characteristic
is provided which will be adequate for many purposes.
The tool assesses the risetime rate of any specified transient, and
extrapolates between the specified points. Other characteristics which can be set include
the arc voltage, which is the voltage across the gas tube in arc mode, and the holdover
voltage. The latter is the voltage which appears across the gap after the current falls to
a value too low to support an arc.
Consideration of the circuit shown previously, and the nature of the
switching characteristics of the gas discharge tube and the diode, shows that there are
several distinct circuit conditions which may exist. For instance, the plot below from the
tool shows the load voltage and gas discharge tube voltage in response to a 500 volt peak
pulse, of risetime 8 microseconds and decay time 30 microseconds. The load resistance was
50 ohms, the source resistance 2 ohms, and the series elements were 5 ohms and 1mH. The
diode switching voltage was 34 volts.
The plot illustrates the complex behaviour of the hybrid circuit. The blue curve
shows the voltage across the gas discharge tube. The voltage rises up to the calculated
overshoot value of approaching 500 volts, and then collapses rapidly to the arc voltage of
20 volts. The arc voltage is maintained until there is insufficient current to maintain
the arc, after which the voltage at the gas tube rises again, and then decays.
The load voltage (which is the same as the voltage across the
suppression diode), rises initially until the diode clamping voltage is reached. After the
gas tube enters arc mode, sufficient voltage is maintained at the diode due to energy
stored in the inductor to keep the diode switched on for a while, after which the load
voltage falls to below the arc voltage.
When the gas tube switches off, the diode again clamps the load
voltage, until there is insufficient voltage for it to conduct, when the load voltage
finally decays exponentially to zero.
If the peak transient voltage were decreased, there would eventually be
insufficient voltage to switch the gas discharge tube. In this condition, the transient
suppression diode would have to be capable of withstanding the pulse current for the
entire transient duration.
Designs using a hybrid device must initially consider the maximum
transient voltage allowed at the load to arrive at the diode clamping voltage. D.C. or low
frequency considerations will determine the maximum value of series resistance; from the
protection circuit viewpoint, the bigger the value of series resistance, the better. Once
the diode voltage and series resistance are estimated, the specified transient can be
applied, and peak power and current ratings of the diode and gas tube arrived at using the
of the thermal noise in a resistor
Although emc is often concerned with minimising the effects of
externally generated noise upon a system, it should be remembered that all resistive
components produce a wide band non-periodic noise voltage, arising from thermal movement
of electrons within the resistance. The open-circuit rms noise voltage produced by a
resistance is given by:
Vn = ( 4 * k * T * B *
R ) 0.5 volts
where k is Boltzmann's constant ( 1.38 * 10 -23 J / oK), T is the
absolute temperature (oK), B is the system noise bandwidth (Hz), and R is the
resistance (ohms). The thermal noise can equally be expressed as a current generator.
The magnitude of the thermal noise is independant of whether the resistor
is carbon, or wirewound etc., and depends only on temperature, the magnitude of the
resistance, and the system bandwidth. Intrinsic noise within a resistor is also often
called Johnson noise, after its discoverer.
The bandwidth dependance is regardless of the actual frequency, and the
noise in a 500Hz bandwidth is the same between 1kHz and 1.5kHz, as it is between 1MHz and
The Thermal Noise Tool allows a spot analysis to be made, and up to
5 sets of data to be plotted, as dBuV versus resistance (log scale) plots. For constant
bandwidth, the dBuVolt versus resistance graphs are straight line plots, with noise
increasing with increasing frequency. A typical value for a 100kohm resistance at room
temperature, in a 20kHz bandwidth system would be 15.18dBuV, or 5.74uVolts.
In addition, the Tool calculates the maximum power that can be
transferred from the resistance. This is independant of the actual resistance value, and
is given by:
Pn = k * T * B watts
At 25oC, the available noise power is equal to 4.11 * 10 -21
The other important consideration regarding thermal noise is the peak
value of the noise signal. The calculation on the previous page provides the rms value,
and not the peak value. In fact, the peak noise voltage probability density follows a
Gaussian distribution, as shown below.
As the distribution never reaches zero, there is a finite chance of
extremely high noise voltages. However as indicated in the diagram, the chance of the peak
voltage exceeding 5 times the rms value is exceedingly small.
Finally it should be noted that only components with a resistive,
energy dissipating component can produce thermal noise. A capacitor for instance cannot
generate thermal noise. The noise generating value of an arbitrary network of passive
components is equal to the real part of the equivalent impedance of the network.